A text teaches you a variety of 5V to 3.3V level application circuit design

5V→3.3V active clamp

One problem with using diode clamps is that it will inject current into the 3.3V supply. In designs with high current 5V output and lightly loaded 3.3V rails, this current injection may cause the 3.3V supply voltage to exceed 3.3V. To avoid this problem, a triode can be used instead, and the triode causes excess output drive current to flow to ground instead of the 3.3V supply. The designed circuit is shown in Figure 11-1.

The base-emitter junction of Q1 acts the same as the diode in the diode clamp circuit. The difference is that only a few percent of the emitter current flows out of the base into the 3.3V rail, and most of the current flows to the collector, and then flows harmlessly from the collector into the ground. The ratio of the base current to the collector current is determined by the current gain of the transistor, typically 10-400, depending on the transistor used.

5V→3.3V resistor divider

A simple resistor divider can be used to reduce the output of a 5V device to a level suitable for a 3.3V device input. The equivalent circuit of this interface is shown in Figure 12-1.

In general, the source resistance RS is very small (less than 10 Ω), and if the selected R1 is much larger than RS, then the effect of RS on R1 can be ignored. At the receiving end, the load resistance RL is very large (greater than 500 kΩ). If the selected R2 is much smaller than RL, then the effect of RL on R2 can be ignored.

There is a trade-off between power consumption and transient time. In order to minimize the power requirements of the interface current, the series resistors R1 and R2 should be as large as possible. However, the load capacitance (composed by the stray capacitance CS and the input capacitance CL of the 3.3V device) can adversely affect the rise and fall times of the input signal. If R1 and R2 are too large, the rise and fall times may be too long to be acceptable.

If the effects of RS and RL are ignored, then the equations for determining R1 and R2 are given by Equation 12-1 below.

Equation 12-2 gives the formula for determining the rise and fall times. To facilitate circuit analysis, the Thevenin equivalent calculation is used to determine the applied voltage VA and the series resistance R. The Thevenin equivalent calculation is defined as the open circuit voltage divided by the short circuit current. According to the limits imposed by Equation 12-2, for the circuit shown in Figure 12-1, the determined Thevenin equivalent resistance R should be 0.66*R1 and the Thevenin equivalent voltage VA should be 0.66*VS.

For example, suppose the following conditions exist:

• Stray capacitance = 30 pF

• Load capacitance = 5 pF

• Maximum rise time from 0.3V to 3V ≤ 1 μs

• External source voltage Vs = 5V

The calculation of the maximum resistance is shown in Equation 12-3.

3.3V→5V level shifter

Although level shifting can be done discretely, it is often more popular to use an integrated solution. Level shifters are used in a wide range of applications: one-way and two-way configurations, different voltage transitions and different speeds for the user to choose the best solution.

Board-level communication between devices (for example, MCU to peripherals) is done via SPI or I2CTM, which is the most common. For SPI, a unidirectional level shifter is appropriate; for I2C, a two-way solution is required. Figure 13-1 below shows these two solutions.

simulation

The final challenge of the 3.3V to 5V interface is how to convert the analog signal across the power barrier. Low-level signals may not require external circuitry, but systems that transmit signals between 3.3V and 5V are subject to power supply variations. For example, in a 3.3V system, the ADC converts a 1V peak analog signal with a higher resolution than the ADC conversion in a 5V system because more of the ADC range is used for conversion in a 3.3V ADC. On the other hand, the relatively high signal amplitude in a 3.3V system may conflict with the lower common-mode voltage limit of the system.

Therefore, in order to compensate for the above differences, some kind of interface circuit may be required. This section discusses interface circuits to help alleviate the problem of signal transitions between different power supplies.

3.3V→5V analog gain module

When connecting from a 3.3V power supply to 5V, the analog voltage needs to be boosted. The 33 kΩ and 17kΩ resistors set the gain of the op amp so that full scale is used at both ends. The 11 kΩ resistor limits the current flowing back to the 3.3V circuit.

3.3V→5V analog compensation module

This module is used to compensate for analog voltages from 3.3V to 5V. The following is the conversion of the analog voltage supplied from a 3.3V power supply to a 5V power supply. The 147 kΩ, 30.1 kΩ resistor on the top right and the +5V supply are equivalent to a 0.85V supply with a 25 kΩ resistor in series. This equivalent 25 kΩ resistor, three 25 kΩ resistors, and an op amp form a differential amplifier with a gain of 1 V/V. The 0.85V equivalent voltage source translates any signal present at the input up the same amplitude up; the signal centered at 3.3V/2 = 1.65V will be centered at 5.0V/2 = 2.50V. The resistor on the upper left limits the current from the 5V circuit.

5V→3.3V active analog attenuator

This technique uses an op amp to attenuate the signal amplitude from a 5V to 3.3V system.

The easiest way to convert a 5V analog signal to a 3.3V analog signal is to use a resistor divider with a R1:R2 ratio of 1.7:3.3. However, there are some problems with this approach.

1) The attenuator may be connected to a capacitive load to form an undesired low pass filter.

2) The attenuator circuit may need to drive a low impedance load from a high impedance source.

In either case, an op amp is needed to buffer the signal.

The desired op amp circuit is a unity gain follower (see Figure 16-1).

The circuit output voltage is the same as the voltage applied to the input.

In order to convert the 5V signal to a lower 3V signal, we only need to add a resistor attenuator.

If the resistor divider is placed before the unity gain follower, it will provide the lowest impedance for the 3.3V circuit. In addition, the op amp can be powered from 3.3V, which will save some power. If the selected X is very large, the power consumption on the 5V side can be minimized.

If the attenuator is behind a unity gain follower, then there is the highest impedance for the 5V source. The op amp must be powered from 5V and the impedance on the 3V side will depend on the value of R1||R2.

5V→3.3V analog limiter

When transferring a 5V signal to a 3.3V system, the attenuation can sometimes be used as a gain. If the desired signal is less than 5V, then sending the signal directly to the 3.3V ADC will result in a larger conversion value. Danger occurs when the signal is close to 5V. Therefore, there is a need to control the voltage overshoot without affecting the voltage in the normal range. Three implementation methods are discussed here.

1. Using a diode, clamp the overvoltage to a 3.3V supply system.

2. Use a Zener diode to clamp the voltage to any desired voltage limit.

3. Use an op amp with a diode for precise clamping.

The easiest way to perform voltage clamping is the same as the simple method of connecting a 5V digital signal to a 3.3V digital signal. Use resistors and diodes to allow excess current to flow into the 3.3V supply. The resistor value chosen must protect the diode and 3.3V supply without negatively impacting analog performance. If the impedance of the 3.3V supply is too low, this type of clamp can cause the 3.3V supply voltage to rise. Even if the 3.3V supply has a very good low impedance, when the diode is on, and at a high enough frequency, when the diode is not conducting (due to parasitic capacitance across the diode), such a clamp will make the input The signal applies noise to the 3.3V supply.

In order to prevent the input signal from affecting the power supply, or to make the input more comfortable when dealing with large transient currents, the above method is slightly changed, and the Zener diode is used instead. Zener diodes are typically slower than the fast signal diodes used in the first circuit. However, Zener clamps are generally more robust, and clamps do not depend on the characteristics of the power supply. The size of the clamp depends on the current flowing through the diode. This is determined by the value of R1. R1 is not required if the output impedance of the VIN source is large enough.

If you need a more accurate overvoltage clamp that does not depend on the power supply, you can use an op amp to get a precision diode. The circuit is shown in Figure 17-3. The op amp compensates for the forward voltage drop of the diode so that the voltage is clamped to the supply voltage of the non-inverting input of the op amp. If the op amp is rail-to-rail, it can be powered from 3.3V.

Since the clamp is performed by an op amp, it does not affect the power supply.

The op amp does not improve the impedance present in the low voltage circuit, and the impedance is still R1 plus the source circuit impedance.

Driving bipolar transistor

When driving a bipolar transistor, the base "drive" current and forward current gain (Î’/hFE) will determine how much current the transistor will sink. If the transistor is driven by the microcontroller I/O port, the base voltage is calculated using the port voltage and the upper port current (typically 20 mA). If 3.3V technology is used, the base current limiting resistor with a lower value should be used to ensure that there is enough base drive current to saturate the transistor.

The value of RBASE depends on the microcontroller supply voltage. Equation 18-1 shows how to calculate RBASE.

If a bipolar transistor is used as a switch to turn the load controlled by the microcontroller I/O port pins on or off, the minimum hFE specification and margin should be used to ensure full saturation of the device.

3V technology example:

For these two examples, it is good practice to increase the base current margin. Driving a 1mA base current to 2 mA ensures saturation, but at the expense of increased input power.

Driving N-Channel MOSFET Transistor

Care must be taken when selecting an external N-channel MOSFET for use with a 3.3V microcontroller. The MOSFET gate threshold voltage indicates the ability of the device to fully saturate. For 3.3V applications, the rated on-resistance of the selected MOSFET should be for a gate drive voltage of 3V or less. For example, for a 100 mA load with a 3.3V drive, a FET with a nominal drain current of 250 μA does not necessarily provide satisfactory results when the gate-source is applied with 1V. When switching from 5V to 3V, the gate-source threshold and on-resistance characteristics should be carefully examined, as shown in Figure 19-1. A slight reduction in the gate drive voltage can significantly reduce the leakage current.

For MOSFETs, low-threshold devices are more common and have a drain-to-source voltage rating below 30V. MOSFETs with a drain-source rated voltage greater than 30V typically have a higher threshold voltage (VT).

As shown in Table 19-1, the threshold voltage of this 30V N-channel MOSFET switch is 0.6V. The MOSFET's rated resistance is 35 mΩ when the gate is applied with 2.8V, making it ideal for 3.3V applications.

For the specifications in the IRF7201 data sheet, the minimum gate threshold voltage is specified as 1.0V. This does not mean that the device can be used to switch current at 1.0V gate-to-source voltage, as there is no specification for VGS (th) below 4.5V. For 3.3V drives requiring low switching resistance, the IRF7201 is not recommended, but it can be used in 5V drive applications.